Current-sensing amplifiers output a voltage proportional to an input current. They may use a resistor to convert the input current into a corresponding voltage, which is then amplified. Among many other applications, current-sensing amplifiers may be used for motor and solenoid control in, for example, automotive power-steering and adaptive-suspension systems, industrial-process control, and medical applications. FIGS. 1A and 1B illustrate diagrams of solenoid 100 and motor control 150 applications, respectively. In such applications, the input 102 receives a pulse-width modulated waveform 104 that toggles between (for example) −2 V and 24 V, 48 V, or 80 V. The rising and falling time of the transitions of the input 102 may be as small as approximately 10 nS, and the frequency of the input 102 may be as great as approximately 20 kHz. A current-sensing amplifier 106 in such applications may be judged on the offset (i.e., a difference between real and expected output values), drift (i.e., a change in output values despite constant input values), and common-mode step response of its output. Ideally, the amplifier 106 produces a result based on the difference between its inputs, regardless of the actual values of the inputs (i.e., their common-mode level); in practice, however, the output of the amplifier 106 may change at different common-mode levels of its inputs. For example, if the amplifier 106 is tuned to remove an offset at a first common-mode level, the tuning may need to be adjusted at a second common-mode level to remove a new offset introduced by the new common-mode level.
The common-mode step response of the amplifier 106 may be especially important in applications having large changes in the input common mode voltage; while the amplifier 106 is recovering from the change in input common mode voltage, the output of the amplifier may not be valid due to the new offset induced by the new common-mode level. Thus, a long settling time of the amplifier 106 (and thus the large error during that period of time) may seriously degrade the dynamic performance of the amplifier 106. In addition, such amplifiers typically have an unacceptably large DC offset, offset drift, and poor CMRR, thus making them unsuitable for precision applications.
In order to improve the DC precision of the amplifier, an auto-zero technique may be used. FIG. 2 illustrates an example of a “ping-pong” auto-zero amplifier 200. It has two input paths 202, 204 disposed in parallel: each path 202, 204 includes a main amplifier 206, an auxiliary differential pair 208 to correct the offset of the main amplifier, and a pair 210 of offset-storage capacitors. The offset-storage capacitors 210 sample the voltage on the outputs 212 of the main amplifiers 206 and feed the samples back, via the auxiliary differential pairs 208, to tuning the main amplifiers 206 to correct any offset therein. Each path 202, 204 may be calibrated periodically and alternatively, in accordance with, for example, an auto-zero clock, so that the offset-correction voltage is refreshed periodically. In other words, while one path 202 is amplifying the input signal, the other path is calibrating itself, and vice-versa. Such an auto-zero amplifier 200 may achieve very low DC offset, offset drift, and high DC CMRR.
It may take a relatively long time, however, for a traditional auto-zero amplifier to recover (i.e., cancel a new offset) after a step in the input common-mode voltage and, during recovery, the output of the amplifier may be invalid. An offset in an amplifier may result from a mismatch between devices' transconductance and/or mismatching between devices' output impedance. The degree of mismatching may be affected by device bias current, MOSFET drain-to-source voltage and back-gate bias voltage, and/or bipolar transistor collector-to-emitter voltage. All of these factors may be affected by input common mode voltage. Because the amplifier typically has different offsets at different common-mode voltages, and thus requires different offset-correction voltages to correct these offsets, this long recovery time hinders the accuracy of the amplifier. The recovery time varies significantly: it may depend on the unpredictable timing relationship between the auto-zero cycle and input common mode step and/or the auto-zero clock frequency, which varies with temperature and process corner. In other words, if a sudden step in the input common-mode voltage occurs at a first time t0 and creates an offset in the output of the auto-zero amplifier, it may not be corrected until a later time t1 during the next auto-zero cycle. The times t0 and t1, and the length of time between them, may be unknown and unpredictable. Although the auto-zero frequency may be increased to reduce the length of such time, nevertheless, due to the discrete nature of auto-zero operation, the amplifier is still unable to start the settling process immediately after a common mode input step. Furthermore, in practice, the settling time of the auto-zero calibration loop, power consumption, switch charge injection, etc., may limit how fast the auto-zero frequency can be. Other techniques used to improve the DC accuracy of a current-sensing amplifier, such as chopper stabilization, have the same drawbacks due to, for example, the long settling time of a capacitor in an internal filter.
FIG. 3 illustrates a common-mode step response 300 of a traditional ping-pong auto-zero amplifier as a function of time 302 (in microseconds). An input common-mode voltage 304 toggles between a wide common-mode range (in this example, between −2 V 306 and 80 V 308, but the invention is not limited to these values); an auto-zero control logic signal 310 toggles between high 312 and low 314 values to enable and disable two paths of a ping-pong amplifier (such as the paths 202, 204 described above with reference to FIG. 2). When the control signal 310 is at logic high 312, a first ping-pong main amplifier is in auto-zero mode and the a second ping-pong main amplifier is in the signal path; conversely, when the control signal 310 is at logic low 314, the second ping-pong main amplifier is in auto-zero mode and the first ping-pong main amplifier is in the signal path. The output 300 of the overall amplifier is, in this example, configured to have a gain of 20.
As is shown in FIG. 3, the settling times 316, 318 of the output 300 in response to the transitions in the common-mode input 304 may be long and unpredictable. For example, at approximately 29 uS, the common-mode voltage 304 steps from 80 V down to −2V, and a large offset appears at the amplifier output 300; this offset persists until approximately 35 uS. The offset persists for so long because of the timing of the input step 304 and the auto-zero control 310. Between 29 uS and 30 uS, the first ping-pong main amplifier is in the signal path; its offset-correction voltage was calibrated, during its auto-zero mode, at a common-mode voltage of 80 V. Between 26.5 uS and 30.5 uS, the second main amplifier is in auto-zero mode. At 29 uS, however, the common-mode input 304 suddenly changes, which also changes the offset of the first and second main amplifiers. Between 29 uS and 30.5 uS, a very large error 320 (e.g., >150 mV) appears at the output 300 due to the sudden offset change of the first main amplifier resulting from the large change in the common-mode level of the input 304. During the same time, the second main amplifier also tries to find the new offset-correction voltage for the new common mode level; it has only 1.5 uS to do so, however. The second main amplifier does thus not have enough time to settle to the correct offset-correction voltage by the end of its auto-zero cycle. At 30.5 uS, the second main amplifier switches from auto-zero mode into the signal path, producing an undesirable offset 322 (e.g. a bump riding on the output 300) between approximately 30.5 uS and 35 uS. The output 300 finally settles at its correct value at 35 uS, after the first main amplifier is in auto-zero mode at −2V between 30.5 uS to 35 uS and then switches back into the signal path. The total common mode step recovery time is thus 6 uS. The next common-mode step happens at 54 uS; in this case, the overall recovery time is about 4 uS for similar reasons.
Thus, existing current-sensing amplifiers may not properly handle large steps in input common mode voltage. A need therefore exists for a cost-effective and precise way to compensate for large and/or fast changes in an input common-mode voltage.